A function of automatically adjusting an amplification gain of an amplifier is referred to as an AGC. Further, a type of the AGC which adjusts a gain by varying the amplification gain in a stepped manner (discretely) is referred to as a stepped AGC. In some cases, the stepped AGC is applied in an OFDM-modulated digital broadcasting reception device for receiving digital broadcasting based on standards such as an ISDB-T (Integrated Service Digital Broadcasting Terrestrial) and a DVB-T (Digital Video Broadcasting Terrestrial).
FIG. 8 is a diagram illustrating an exemplary structure of a conventional OFDM-modulated digital broadcasting reception device 300 that performs the stepped AGC (hereinafter referred to as a reception device 300). As shown in FIG. 8, the reception device 300 comprises an antenna 301, a tuner 302, an analog-digital converter (hereinafter referred to as an ADC) 303, a quadrature demodulation section 304, a level detection section 305, a timing control section 306, a synchronization section 307, an FFT section 308, an equalization section 309, an error correction section 310, a data decoding section 311, and a display section 312.
Hereinafter, an operation of the reception device 300 will be described. The antenna 301 receives an OFDM modulated signal so as to output the received OFDM modulated signal to the tuner 302. The tuner 302 selects a signal having a desired channel from the inputted OFDM modulated signal, converts the selected signal into a predetermined intermediate frequency signal (having a center frequency of 57 MHz, for example) so as to amplify the converted intermediate frequency signal, and finally outputs the signal thus obtained to the ADC 303. Note that the tuner 302 includes a gain variable amplifier for amplifying an intermediate frequency signal. FIG. 9 is a diagram illustrating a relationship between a control signal and a gain of the gain variable amplifier included in the tuner 302. As shown in FIG. 9, a gain of the gain variable amplifier discretely varies because a control signal having discrete values generated by digital signal processing (an AGC control signal) is inputted to the gain variable amplifier. Furthermore, the gain of the gain variable amplifier is increased in increments of 1 dB, for example, so as to correspond to a minimum increment of the control signal. The ADC 303 converts the inputted intermediate frequency signal into a digital signal from an analog signal, and outputs the converted signal to the quadrature demodulation section 304. The quadrature demodulation section 304 performs orthogonal detection on the inputted intermediate frequency signal so as to be converted into an I/Q signal, and outputs the I/Q signal thus obtained to the level detection section 305, the synchronization section 307 and the FFT section 308.
The level detection section 305 detects a difference between an electric power level of the inputted I/Q signal and a desired electric power level, and generates an AGC control signal, which is a control signal to be inputted to the gain variable amplifier of the tuner 302. The synchronization section 307 detects an FFT window position which is a position, of the I/Q signal, at which an FFT (Fast Fourier Transform) is performed by using a correlation with respect to a guard period of the inputted I/Q signal, and notifies the FFT section 308 and the timing control section 306 of the detected FFT window position. The FFT section 308 performs the FFT at the notified FFT window position of the I/Q signal so as to convert the I/Q signal into a signal in a frequency range, and outputs the obtained signal to the equalization section 309. The timing control section 306 calculates a timing at which a stepped AGC control signal is to be outputted to the tuner 302 by using information on the notified FFT window position, and outputs the stepped AGC control signal inputted from the level detection section 305 to the tuner 302 at the timing having been calculated. The gain variable amplifier of the tuner 302 varies an amplification gain in accordance with the stepped AGC control signal inputted thereto.
The equalization section 309 executes an equalization process by performing time axis interpolation and frequency axis interpolation, both of which are to be described in detail below, on the inputted I/Q signal in a frequency range, and then outputs the obtained signal to the error correction section 310. Thus, the equalization section 309 is able to estimate time-based and frequency-based errors generated by the OFDM modulated signal being propagated via a transmission path (space) between a transmission device (not shown) and the reception device 300 (hereinafter referred to as transmission path fluctuation) and to correct the estimated errors. The error correction section 310 performs deinterleave, Viterbi decoding and RS (Read-Solomon) decoding, all of which processing are an error correction process, on the corrected I/Q signal, and outputs a TS (Transport Stream) signal thus obtained to the data decoding section 311. The data decoding section 311 executes a video and an audio data decompression process on the inputted TS signal in compliance with the MPEG2 standard or the like, and outputs the TS signal on which the aforementioned processes have been executed to the display section 312. By using the inputted signal, the display section 312 provides a user with services relating to video or the like.
Hereinafter, the level detection section 305, the synchronization section 307 and the timing control section 306 will be more specifically described.
FIG. 10 is a diagram illustrating an exemplary structure of the level detection section 305. As shown in FIG. 10, the level detection section 305 includes an electric power calculation section 330, a logarithmic conversion section 331, a subtraction section 332, a reference value generation section 333, a loop filter section 334, and a quantization section 335. The electric power calculation section 330 calculates I2+Q2 using the I/Q signal inputted from the quadrature demodulation section 304 so as to obtain a received electric power, and outputs the result to the logarithmic conversion section 331. Note that in the case where the AGC is performed with reference to a mean value along a time axis of the received electric power, the electric power calculation section 330 further averages the calculated electric power along the time axis thereof. In the case where the AGC is performed with reference to a peak of the received electric power, the electric power calculation section 330 performs peak detection. The logarithmic conversion section 331 executes a process so as to represent an output of the electric power calculation section 330 (the calculated electric power) using logarithms, and outputs the obtained result to the subtraction section 332. This is because a gain of the variable gain amplifier included in the tuner 302 varies in increments of a predetermined dB value (logarithmic value) with respect to the AGC control signal inputted thereto, as already described above with reference to FIG. 9. The reference value generation section 333 generates a reference electric power for causing an output electric power of the variable gain amplifier included in the tuner 302 to converge to a target value, and outputs the generated reference electric power to the subtraction section 332. The subtraction section 332 detects a difference between an output electric power of the logarithmic conversion section 331 and the reference electric power of the reference value generation section 333, and outputs the detected difference to the loop filter section 334. The loop filter section 334 integrates the inputted difference, and outputs the obtained integral value to the quantization section 335. The quantization section 335 quantizes the integral value inputted thereto, and outputs the quantized value to the timing control section 306 as an AGC control signal. Note that the AGC control signal varied in a minimum increment corresponds to a gain, of the variable gain amplifier of the tuner 302, which varies in increments of 1 dB, for example.
FIG. 11 is a diagram describing an operation of the synchronization section 307. To the synchronization section 307, an I/Q signal (a to-be-demodulated signal) outputted from the quadrature demodulation section 304 is inputted (see (a)). As shown in (a), guard periods are indicated by diagonally shaded portions, and all other portions indicate useful symbol periods. A symbol period is constituted by a guard period and a useful symbol period being arranged as a pair. A guard period is constituted by a signal obtained by replicating a signal existing in a trailing end portion of the useful symbol period with which the guard period is paired up. Then, the synchronization section 307 generates a delay signal obtained by delaying the to-be-demodulated signal (a) by the useful symbol period (see (b)). Thereafter, the synchronization section 307 generates a correlation signal of (a) the to-be-demodulated signal (a) and the delay signal (b) (see (c)). As shown in the correlation signal (c), a correlation appears for a period corresponding to the trailing end portion of the useful symbol period of the to-be-demodulated signal (a). Then, the synchronization section 307 performs shift integration on the correlation signal (c) using a guard period length, thereby generating a shift integral signal (see (d)). The shift integral signal (d) reaches its peak at each boundary (see arrows) of the symbol period of the to-be-demodulated signal (a). Next, the synchronization section 307 detects a boundary of each symbol period (a boundary of each OFDM symbol) of the to-be-demodulated signal (a) by using the peak of the shift integral signal (d). Thereafter, in a fundamental environment (non-multipath environment) where neither delay wave nor preceding wave exists, the synchronization section 307 sets, as the FFT window position at which the FFT is performed, a period B obtained by removing a period A corresponding to the guard period length from said each symbol period, the period A being arranged adjacent to another period A such that the two continuously-arranged periods A have their center at the detected boundary of said each symbol period (see (e)). Note that in a reception environment where a delay wave having a guard period length exists (delay wave environment), the synchronization section 307 sets, as the FFT window position, a position delayed, from the FFT window position set in the fundamental wave environment, by a half of the period corresponding to the guard period length (see (f)). Furthermore, in a reception environment where a preceding wave having a guard period length exists (preceding wave environment), the synchronization section 307 sets, as the FFT window position, a position preceding, the FFT window position set in the fundamental wave environment, by a half of the period corresponding to the guard period length (see (g)). Then, the synchronization section 307 notifies the FFT section 308 and the timing control section 306 of the FFT window positions having been set. The FFT section 308 performs the FFT process on the inputted I/Q signal at each FFT window position (in the period B) having been notified.
Based on said each notified FFT window position (the period B), the timing control section 306 calculates the period A in which no FFT process is performed. Next, the timing control section 306 outputs, within the calculated period A, an AGC control signal inputted from the level detection section 305 to the variable gain amplifier of the tuner 302. Thus, the variable gain amplifier of the tuner 302 is able to vary a gain of an OFDM signal in a stepped manner (discretely) within the period A in which no FFT is performed on the OFDM signal (see (e) to (g) shown in FIG. 11).
As described above, in the case where the stepped AGC is performed, the communication device 300 detects a period (FFT window position) from which a period in which the FFT is performed on an OFDM signal is removed, thereby varying a gain of the OFDM signal in a stepped manner (discretely) within the period in which no FFT is performed. Thus, the communication device 300 is not to be influenced by a discrete gain fluctuation caused by the stepped AGC.
However, the communication device 300 has a problem in that an interpolation error, which is to be described below, occurs in the equalization process executed by the equalization section 309. FIG. 12 is a diagram describing the equalization process executed by the equalization section 309. FIG. 12(b) is an image in which N−6th to Nth symbols of a received OFDM signal are chronologically represented. Note that an OFDM signal is transmitted for each symbol. In the FIG. 12(b), subcarriers included in each symbol are respectively represented by circles, and Mth to M+15th subcarriers are shown as an example of the subcarriers included in each symbol. The Mth to M+15th subcarriers are arranged in an order of modulation frequency. Note that each solid white circle represents a data carrier which is a subcarrier including transmission data. Also, each solid black circle represents a SP (Scattered Pilot) carrier (also referred to as a reference subcarrier) which is a subcarrier whose amplitude and phase at a time of transmission have already been recognized at a reception side. The SP carrier is arranged every 12 carriers in each symbol, and is arranged at a position shifted, from an arrangement position of another SP carrier of an immediately preceding symbol, by three carriers in a direction in which the modulation frequency increases. FIG. 12(a) indicates that a stepped AGC control is performed within a period, in which no FFT is performed), existing between an N−3th symbol and an N−2th symbol (the period A shown in FIG. 11), thereby causing a gain of the variable gain amplifier of the tuner 302 to be multiplied by X.
Hereinafter, a case where the equalization section 309 executes the equalization process on the N−3th symbol will be described as an example. Note that at a time when the equalization process is executed on the N−3th symbol, the N−2th to Nth symbols have already been received and inputted to the equalization section 309. The equalization section 309 compares an amplitude and a phase of an already received SP carrier (herein after referred to as a received SP carrier) with an amplitude and a phase of a SP carrier, at a time of transmission, corresponding to the received SP carrier (hereinafter referred to as a transmitted SP carrier). That is, the equalization section 309 detects distortion of the received SP carrier. Note that the amplitude and the phase of the transmitted SP carrier are already recognized by the equalization section 309. This comparison is made on the respective received SP carriers included in the N−6th to Nth symbols, whereby the equalization section 309 calculates transmission path coefficients, respectively, at positions of the received SP carriers included in the N−6th to Nth symbols. Note that the transmission path coefficient is a value indicating a variation amount of an amplitude and a phase of a received SP carrier with respect to those of a transmitted SP carrier. In other words, a transmission path coefficient is a value indicating distortion of an OFDM signal being transmitted.
Next, the equalization section 309 calculates a transmission path coefficient at a position of each data carrier represented by a solid white circle with a heavy outline included in the N−3th symbol by means of linear interpolation. Note that a transmission path coefficient at a position of an Mth subcarrier included in the Nth symbol is defined as H(N, M).
Firstly, it is assumed that no stepped AGC control is performed, i.e., a gain does not vary at all. In this case, the equalization section 309 calculates a transmission path coefficient H(N−3, M+3) at the position of the M+3th subcarrier (represented by a solid white circle with a heavy outline) included in the N−3th symbol by using a formula 1, which indicates a linear interpolation (time axis interpolation) using a transmission path coefficient H(N−6, M+3) at a position of a solid black circle and a transmission path coefficient H(N−2, M+3) at a position of another solid black circle. Note that the transmission path coefficient H(N−6, M+3) and the transmission path coefficient H(N−2, M−3) are already calculated as described above.H(N−3,M+3)=(¼)*H(N−6,M+3)+(¾)*H(N−2,M+3)  [Formula 1]Similarly, the equalization section 309 performs the time axis interpolation, thereby calculating transmission path coefficients H(N−3, M+6), H(N−3, M+9) and H(N−3, M+15). Note that transmission path coefficients H(N−3, M) and H(N−3, M+12) at the positions of the received SP carriers are already calculated. As a result, the equalization section 309 can calculate a transmission path coefficient at a position of every third subcarrier of all the subcarriers included in the N−3th symbol, starting from the Mth SP carrier. Next, the equalization section 309 performs, on the N−3th symbol, interpolation in a direction along a frequency axis of the subcarriers (frequency axis interpolation), thereby calculating the transmission path coefficients at positions of all the subcarriers included in the N−3th symbol. Then, by using the transmission path coefficients of the positions of all the subcarriers, the equalization section 309 cancels (compensates for) the distortion (amplitude and phase errors) of the OFDM signal being propagated via a transmission path, so as to correct the distortion of the OFDM signal.
Next, it is assumed, as shown in FIG. 12, that the stepped AGC control is performed and thus a gain variation is multiplied by X between the N−3th symbol and the N−2th symbol. In this case, a transmission path coefficient H(N−2, M+3) at a position represented by a solid black circle is multiplied by X due to a stepped gain variation (discrete gain variation) so as to become H(N−2, M+3)*X. Therefore, the equalization section 309 calculates the transmission path coefficient H(N−3, M+3) by using a formula 2 indicating the time axis interpolation performed between symbols along which the gain is varied.H(N−3,M+3)=(¼)*H(N−6,M+3)+(¾)*H(N−2,M+30)*X  [Formula 2]
As indicated above, H(N−3, M+3) of the formula 2 includes an error generated by multiplying H(N−2, M+3) by X with respect to H(N−3, M+3) of formula 1. Similarly, the equalization section 309 performs the time axis interpolation between the symbols along which a gain is varied, thereby calculating the transmission path coefficients H(N−3, M+6), H(N−3, M+9) and H(N−3, M+15), each of the coefficients including an error. Then, the equalization section 309 performs the frequency axis interpolation on the N−3th symbol, thereby calculating the transmission path coefficients at the positions of all the subcarriers included in the N−3th symbol, each of the transmission path coefficients including an error. FIG. 12(c) is an image illustrating how errors generated in the respective symbols shown in FIG. 12(b) (hereinafter referred to as interpolation errors) vary on a symbol-by-symbol basis. In FIG. 12(c), an image enclosed by a rectangular represents an interpolation error of the N−3th symbol. When a heavy line pointed by an arrow indicates a normal transmission path coefficient, an interpolation error is indicated by diagonally shaded areas. Next, by using the transmission path coefficients, each including the interpolation error, the equalization section 309 cancels (compensates for) the distortion (amplitude and phase errors) caused by the OFDM signal being propagated via the transmission path. As a result, a demodulation error of the OFDM signal is generated, thereby resulting in deterioration of the reception performance.
FIG. 13 is a diagram describing a structure of the equalization section 309 included in the reception device 300. As shown in FIG. 13, the equalization section 309 includes a SP separation section 320, a complex division section 321, a SP generation section 322, a time axis interpolation section 324, a frequency axis interpolation section 325, a delay section 326, and a complex division section 327. Hereinafter, operations of the respective components included in the equalization section 309 will be simply described with reference to FIG. 13. To the SP separation section 320, an OFDM signal in a frequency area is inputted from the FFT section 308. The SP separation section 320 separates a SP carrier of the OFDM signal therefrom and outputs the SP carrier to the complex division section 321, and then outputs, to the delay section 326, the OFDM signal from which the SP carrier has been separated. The SP generation section 322 generates a SP carrier having an amplitude and a phase, both of which being synchronized with those of the transmitted SP carrier, and outputs the generated SP carrier to the complex division section 321. The complex division section 321 divides a signal of the received SP carrier separated by the SP separation section 320 by a signal of the transmitted SP carrier generated by the SP generation section 322. Thus, the complex division section 321 calculates a transmission path coefficient at a position of the received SP carrier, and outputs the calculated transmission path coefficient to the time axis interpolation section 324. The time axis interpolation section 324 performs the time axis interpolation which has been described with reference to FIG. 12, and outputs, to the frequency axis interpolation section 325, the transmission path coefficient at the position of the received SP carrier inputted thereto and transmission path coefficients calculated by means of the time axis interpolation. The frequency axis interpolation section 325 performs the frequency axis interpolation, which has been described with reference to FIG. 12, so as to calculate transmission path coefficients at positions of all subcarriers, and outputs the obtained transmission path coefficients to the complex division section 327. The delay section 326 delays the OFDM signal inputted from the SP separation section 320 and outputs the delayed signal to the complex division section 327. The complex division section 327 performs complex division, respectively, on data carriers of the OFDM signal outputted from the delay section 326 by the transmission path coefficients outputted from the frequency axis interpolation section 325, and outputs the obtained OFDM signal to the error correction section 310. By executing the process described above, the equalization section 309 cancels (compensates for) the distortion of the OFDM signal being transmitted via the transmission path. As described with reference to FIG. 12, however, the equalization section 309 generates an interpolation error due to a gain variation caused by the stepped AGC control when the time axis interpolation is performed.
As described above, in the conventional reception device 300, the equalization section 309 malfunctions due to a discrete amplitude fluctuation caused by the stepped AGC, and thus the reception performance deteriorates.
Patent document 1 discloses a technique which solves the problem mentioned above. FIG. 14 is a diagram illustrating an automatic gain control device 400 disclosed in patent document 1. As shown in FIG. 14, the automatic gain control device 400 includes a gain variable section 401, a gain variable section 402, and an automatic gain variable control section 407. The gain variable section 401 provides an inputted modulated signal with stepped gain variation. The gain variable section 402 provides, with linear gain variation, a signal whose amplitude varies in a stepped manner, which is outputted from the gain variable section 401. The automatic gain variable control section 407 controls the gain variable section 401 and the gain variable section 402 so as to maintain an amplitude of the signal outputted from the gain variable section 402 at a constant.
FIG. 15 is a diagram describing amplitudes and gains of input and output signals of the gain variable section 401 and the gain variable section 402. FIG. 15(a) shows the amplitude of a signal inputted to the gain variable section 401. FIG. 15(b) shows a steppedly-varied gain with which the gain variable section 401 provides the signal inputted thereto. FIG. 15(c) shows the amplitude of the signal outputted from the gain variable section 401. FIG. 15(d) shows a linearly-varied gain with which the gain variable section 402 provides the signal inputted thereto. FIG. 15(e) shows the amplitude of the signal outputted from the gain variable section 402. For example, if the amplitude of a signal inputted to the gain variable section 401 gradually decreases (see (a)), the gain variable section 401 provides the inputted signal with a steppedly-varied gain (see (b)). As a result, the gain variable section 401 outputs a signal whose amplitude varies discretely (see (c)). In response to an instruction from the automatic gain variable control section 407, the gain variable section 402 provides a linearly-varied gain with an output signal of the gain variable section 401 so as to maintain an amplitude of the signal at a constant (see (d)). As a result, the gain variable section 402 is able to output a signal whose amplitude is maintained constant (see (e)).
As described above, the automatic gain control device 400 is able to maintain the amplitude of a signal outputted from the gain variable amplifier at a constant even when the stepped AGC is performed. Thus, by utilizing the automatic gain control device 400 in the gain variable amplifier of the tuner 302 included in the reception device 300 (see FIG. 8), it may be possible not to cause the equalization section 309 to malfunction due to the discrete amplitude fluctuation caused by the stepped AGC.    [Patent document 1] Japanese Laid-Open Patent Publication No. 2006-74702